Clock recovery for optical transmission systems

ABSTRACT

A receiver for an optical communications system which corrects distortion of a received signal. A clock recovery system utilizing a feedback and feedforward system are provided. The feedback loop comprises a phase detector and a clock source, while the feedforward loop comprises the phase detector and a delay element for delaying the output of distortion correction system. The feedback loop has a significantly lower bandwidth than the feedforward path. There are also provided methods of optimizing tap weights and of acquiring initial tap weights.

CROSS REFERENCE TO RELATED APPLICATION

This application is a divisional of U.S. application Ser. No.13/322,938, filed Nov. 11, 2011, which is a national phase filing of PCTApplication No. PCT/IB2010/053252, filed Jul. 16, 2010, which claimspriority to UK Patent Application No. 0912881.0, filed Jul. 23, 2009,the entirety of each being incorporated herein by reference.

BACKGROUND

This invention relates to a receiver for an optical communicationssystem, and in particular to clock recovery in a Dual PolarisationQuadrature Phase Shift Keying System.

Two principle forms of modulation are utilised in optical communicationssystems; Amplitude Shift Keying (ASK) and Phase Shift Keying (PSK),which encode data in the amplitude and phase, respectively, of thetransmitted light. Direct detection methods can be utilised to detectand receive ASK signals, but not PSK signals in which the data cannot berecovered from the power envelope of the light. Coherent detection, inwhich the received light is mixed with an optical Local Oscillator (LO),enables the reception of PSK signals.

The optical LO may be locked to the frequency and phase of the incomingoptical signal (homodyne reception), or, may be held very close to, butnot locked precisely to, the incoming optical signal (intradynereception), or may be at a significantly different frequency in relationto the incoming optical signal (heterodyne reception). Locking anoptical LO to an incoming signal for a homodyne system presents manypractical difficulties in the optical implementation, while heterodynereception requires the use of high frequency electronics to remove thefrequency offset. Intradyne reception offers a compromise where controlof the optical LO is relatively easy to achieve, and the bandwidth ofthe electrical signal is kept to frequencies which are also relativelysimple to manage and process.

A particular form of PSK is Quadrature PSK (QPSK) in which two bits areencoded per symbol. The symbol rate of a QPSK signal is thus half thebit rate carried by the signal. FIG. 1 shows power 10 and electricalfield 11 eyes of a QPSK signal. As can be seen, the optical power ofeach symbol is the same, with the information residing in the opticalphase of the signal. Variations in the optical power envelope are causedby transitions between symbols and do not convey any information.

Light sources used for optical transmission systems are generallywell-polarised lasers. Independently modulated sources can thus bepolarisation multiplexed for transmission, thereby transmitting fourbits per symbol at a single wavelength in a Dual Polarisation QPSK(DP-QPSK) format. In a DP-QPSK signal the power and field eyes shown inFIG. 1 are repeated independently, although usually aligned in time atthe transmitter, on each polarisation.

A DP-QPSK optical signal is conveniently generated from 4 independentdata signals, each at the symbol rate. A 40 Gb/s DP-QPSK signal can thusbe generated from four 10 Gb/s electrical signals, thereby utilisingrelatively cheap 10 Gb/s electrical components. FIG. 2 shows a schematicdiagram of a DP-QPSK modulator driven by four signals 20-23, at thesymbol rate.

FIG. 3 shows an example of a receiver for receiving a DP-QPSK signal.The received signal 30 is split into two orthogonal polarisations bypolarisation beam splitter 31 and each signal is fed to a 90° opticalhybrid 32, 33. An optical LO 34 is also fed to each hybrid 32, 33 formixing with the data signals. The outputs of each hybrid are passed toseparate photodetectors 35 a, b, c, d to convert their amplitudes toelectrical signals which are converted to digital values by Analogue toDigital Converters (ADCs) 36 a, b, c, d. Those values are passed to ASIC37 for digital signal processing.

The outputs from the photodiodes can be expressed as shown below.V _(x0) =|E _(sx) +E _(LO)|² =|E _(sx)|² +|E _(LO)|²+2R _(e) {E _(sx) E_(LO)*}V _(x90) =|E _(sx) +jE _(LO)|² =|E _(sx)|² +|E _(LO)|²+2T _(m) {E _(sx)E _(LO)*}V _(y0) =|E _(sy) +E _(LO)|² =|E _(sy)|²+|²+2R _(e) {E _(sy) E _(LO)*}V _(x90) =|E _(sy) +E _(LO)|² =|E _(sy)|² +|E _(LO)|²+2T _(m) {E _(sy) E_(LO)*}The first two terms on the right in each equation are small, or can beremoved by electrical components, leaving the detected signalsrepresented by the right hand term on each line. Each of the electricalsignals passed to ASIC 37 thus represent a combination of the datasignal and the optical LO. The ASIC must therefore remove the residualLO from the signals to enable decoding of the data.

Optical signals suffer distortion during their transmission, for exampledue to chromatic dispersion. It is known that Finite Impulse Response(FIR) filters are effective at removing linear dispersions such asChromatic Dispersion (CD) (see, for example, J. H. Winters,“Equalization in Coherent Lightwave Systems Using a Fractionally SpacedEqualizer”, JLT, Vol. 8, No. 10, October 1990 and Taylor, M. (2004),‘Coherent detection method using DSP for demodulation of signal andsubsequent equalization of propagation impairments’, PhotonicsTechnology Letters, IEEE 16(2), 674-676.), both of which areincorporated herein by reference. FIG. 4 shows a simplified blockdiagram of a receiver for a single polarisation utilizing an FIR filter40 to correct distortion and a carrier recovery block 41 to remove theresidual LO offset signal.

After correction by the FIR filter 40 the symbols are discrete, butlocated at arbitrary phases (as shown at 42) due to the LO offset. Thecarrier recovery block 41 removes that offset resulting in the phaseslying on the expected constellation 43 for a QPSK signal.

FIG. 5 shows an example configuration of a FIR filter 40, implemented asis known in the art.

During transmission the polarisation of the optical signal is rotatedand may be received in any arbitrary alignment, not necessarily alignedwith the receiver as has been assumed above. A butterfly structure ofFIR filters may be utilised to process the received signals when thepolarisation is in an unknown state. FIG. 6 shows a filter structure forperforming this demultiplexing as described in, for example, Savory etal., “Digital Equalisation of 40 Gbit/s per Wavelength Transmission over2480 km of Standard Fibre without Optical Dispersion Compensation”,ECOC2006, Paper 2.5.5, 2006, incorporated herein by reference.

FIG. 7 shows a block diagram of a digital receiver system for an opticalcommunications system. As explained above, the input signal 70 is mixedwith a local oscillator in block 32 and fed to a set of four photodiodes35. The outputs of the photodiodes are digitised in ADCs 36, the outputbeing passed to a digital processing system 71. The digital processingsystem 71 is typically provided by a CMOS Application SpecificIntegrated Circuit (ASIC) specifically designed to process the digitisedsignals including an equaliser 74 to correct distortion, but may be anysystem suitable for performing the tasks required, for example a DSP maybe appropriate. The processing system processes the data in real timeand therefore must be capable of operating on the full data payload. Forexample, a typical receiver may receive a 10-40 Gb/s signal forprocessing. ASICs provide convenient systems for performing thisprocessing as they allow the design of a highly parallel system to copewith processing such high data rates.

The ADCs and processing system and other components may be provided by asingle device, or separated between different devices as appropriate.The same or different type of device may be appropriate for eachfunction.

The data clock frequency and phase of the incoming signal must bederived such that the ADCs can sample the incoming signal at the correctpoint and sample rate. A conventional approach, as shown in FIG. 7, isto use an analogue Phase Locked Loop (PLL) formed of phase detector 73and Voltage Controlled Oscillator (VCO) 72. Although not shownexplicitly, it will be appreciated that a loop filter will beincorporated in the PLL. It is generally convenient to locate this inclose proximity to the VCO and it may therefore be considered to formpart of the VCO block 72 in FIG. 7. However, the incoming signal may beso distorted that this system cannot track the signal to acquire theclock phase. For example, the distortion correction may tolerate10,000-20,000 ps/nm of dispersion which is significantly higher thanconventional analogue PLLs have been shown to work over.

An alternative method shown in FIG. 8 is to utilise a digital phasedetector 80 operating from the equalised signals to control the VCO 72.In order to meet certain telecommunications standards (for exampleG.8251) a PLL bandwidth of greater than 1 MHz may be required. However,the correction system 74 introduces delay into the PLL which affectsoperation. In particular, gain peaking is introduced making it extremelydifficult, if not impossible, to meet the performance required by thestandards. The processor may require 10-40 clock cycles to perform theequalisation process and typically operates at 300-600 MHz giving 30-100ns of delay in the feedback loop which is sufficient to degrade theperformance of a 1 MHz PLL. FIG. 9 shows a Jitter mask demonstrating theeffect of delay on the PLL of FIG. 8 with a 1 MHz bandwidth. Performanceis reduced to below the required level even with 16 clock cycles ofdelay in the loop. In contrast, in similar implementations in the radiodomain the processing rate is far higher than the data rate, making thedelay less significant to the operation of the feedback loop.

There is therefore a requirement for a clock recovery system that canperform clock recovery from a highly distorted signal, such as anuncompensated DP-QPSK signal.

The startup of an optical transmission system receiver may be difficult,or impossible. The clock recovery and compensation systems areinter-dependent and one cannot begin operation without the other beingat least partially operational. There is therefore a need for a methodwhich can initialise the receiver in such a system.

SUMMARY

This Summary is provided to introduce a selection of concepts in asimplified form that are further described below in the DetailedDescription. This Summary is not intended to identify key features oressential features of the claimed subject matter, nor is it intended tobe used as an aid in determining the scope of the claimed subjectmatter.

There is provided a receiver for receiving at least one input signalfrom a photodiode in an optical communications system, comprising anAnalogue to Digital Converter (ADC) configured to digitise the at leastone input signal and output a digitised signal, a digital processingsystem configured to process the digitised signal and output a processedsignal, a phase detector configured to detect the timing phase of theprocessed signal and output a signal indicative of that phase, a clocksource providing a sampling clock signal to the analogue to digitalconverter, the clock source having the signal indicative of the phase ofthe processed signal as an input to control the sampling clock signal,and a delay element configured to delay the processed signal and outputa delayed processed signal, the delay of the delay element beingcontrolled by the output of the phase detector.

The processing of the digital signal may comprise correction ofdistortion introduced in the optical transmission system.

The digital processing system may comprise a Finite Impulse Response(FIR) filter for processing the digitised signal.

The FIR may comprise variable tap weights.

A tap weight update algorithm may be implemented in the processingsystem.

The receiver may further comprise a low pass filter configured to filterthe signal indicative of the phase.

The filtered signal indicative of the phase may be utilised by the delayelement, and the unfiltered signal is utilised by the clock source.

The filtered signal indicative of the phase may be utilised by the delayelement, and the clock source.

The clock source may be implemented in a device with the at least oneADC.

The clock source may be implemented as a separate device to the at leastone ADC.

The at least one ADC and clock source may be implemented in a firstdevice and the processing system is implemented in a second device.

The second device may be an ASIC.

The processing system may process the signals in real time.

The receiver may further comprise at least one photodiode, eachproviding an input to an ADC.

The receiver may comprise four ADCs.

The receiver may further comprise an optical Local Oscillator and atleast one optical hybrid configured to mix the optical Local Oscillatorwith a received optical signal, the output of the at least one opticalhybrid being the input to the at least one photodiode.

The receiver may further comprise a carrier recovery system operating atthe output of the digital processing system.

The receiver may further comprise a decision system configured to decidethe value of received symbols.

The tap update algorithm may utilise inputs comprising a delayed versionof the at least one input signal, the delay being the same as the delayapplied by the delay element to the processed signal, the output of thedelay element, and the decided symbols.

The phase detector may be configured to select one or both of twosignals relating to two polarisations.

The receiver may further comprise a tap weight centralisation system forcentralising the tap weights of the FIR filter.

The receiver may be configured to receive, correct and decode a dualpolarisation quadrature phase shift keyed optical signal.

There is also provided a method of receiving a modulated optical signal,comprising the steps of receiving the optical signal in at least onephotodiode, digitising the output of the photodiode to provide an inputsignal utilising an ADC, processing the input signal to correctdistortion and outputting a corrected signal, monitoring the timingphase of the corrected signal and outputting a phase signal indicativeof that phase, utilising the phase signal to control a clock sourceproviding a clock signal to the ADC, and delaying the corrected signalin accordance with the phase signal.

There is also provided a method of optimising tap weights in an FIRfilter utilised to correct distortion in an optical communicationsreceiver, comprising the steps of monitoring the tap weight centre ofthe FIR filter, calculating the offset of tap weight centre from thecentral tap position of the FIR filter, and utilising that offset todefine a sampling clock phase of an ADC, the output of which is passedto the FIR filter.

The sampling clock phase may be defined by adding a signal indicative ofthe offset to a signal indicative of the phase of a signal output by theFIR filter.

The offset may be utilised to adjust a delay applied to the output ofthe FIR filter.

There is also provided a method of initially acquiring tap weights foran FIR filter used to correct distortion in an optical communicationsreceiver, the method being performed by the receiver and comprising thesteps of acquiring and storing a series of samples of a received signal,and applying a blind optimisation algorithm to the series of samples toobtain an estimate of tap weights for a Finite Impulse Response (FIR)filter of the receiver configured to equalise a received optical signal.

The method may further comprise the step of transferring the series ofsamples to a digital processing system and performing the blindoptimisation in that processing system.

The blind optimisation algorithm may be applied in both forwards andbackwards directions to the series of samples.

The method may further comprise the step of transferring the estimatedtap weights to the FIR filter.

The method may further comprise the step of commencing correctingdistortion in a received signal utilising the FIR filter.

The method may further comprise the step of activating a clock recoverysystem in the receiver.

The method may further comprise the steps of acquiring and storing afurther series of samples of a received signal, and applying a blindoptimisation algorithm to the further series of samples to obtain animproved estimate of the tap weights.

The method may further comprise the step of applying the improved tapweights to the FIR filter and utilising those tap weights for correctionof distortion in a received signal.

The method may further comprise the step of activating a decision systemin the receiver to decode the values represented by a received signal.

The method may further comprise the step of activating a tap updatealgorithm, said algorithm being configured to optimise the tap weights.

The preferred features may be combined as appropriate, as would beapparent to a skilled person, and may be combined with any of theaspects of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the invention will be described, by way of example, withreference to the following drawings, in which:

FIG. 1 shows power and phase eyes for a QPSK signal;

FIG. 2 shows an example QPSK transmitter;

FIG. 3 shows part of a coherent receiver;

FIG. 4 shows a block diagram of a digital receiver for an opticalcommunications system;

FIG. 5 shows an example of an FIR filter;

FIG. 6 shows an example of a receiver for a dual polarisation system;

FIG. 7 shows a block diagram of a receiver using an analogue phase lockloop;

FIG. 8 shows a block diagram of a receiver using a combined digital andanalogue phase lock loop;

FIG. 9 shows a Jitter mask demonstrating the effect of delay on thefeedback loop of FIG. 8 with a 1 MHz bandwidth;

FIG. 10 shows an example of an optical receiver using combined feedbackand feed forward signals for clock recovery;

FIG. 11 shows a jitter mask;

FIG. 12 shows a block diagram of an optical communications receiver;

FIG. 13 shows a phase detecting system;

FIG. 14 shows a graph of the required phase detector output;

FIG. 15 shows a clock lock detecting system;

FIGS. 16A and 16B show examples of tap weights in an FIR filter;

FIG. 17 shows a method of clock acquisition;

FIG. 18 shows an example of clock and tap weight acquisition; and

FIG. 19 shows a different view of the method of FIG. 18.

DETAILED DESCRIPTION

Embodiments of the present invention are described below by way ofexample only. These examples represent the best ways of putting theinvention into practice that are currently known to the Applicantalthough they are not the only ways in which this could be achieved.

The description sets forth the functions of the example and the sequenceof steps for constructing and operating the example. However, the sameor equivalent functions and sequences may be accomplished by differentexamples.

A 40 Gbit/s DP-QPSK transmission system is utilised as the basis for thefollowing description, but the techniques described are applicable to arange of transmission formats and rates without unduly burdensomemodification.

FIG. 10 shows a block diagram of a clock recovery system, using afeedback loop 90 and a feedforward path 91. Phase Detector 92 detectsthe phase of the output of equaliser 74. The phase detector 92 output isfed back to Voltage Controlled Oscillator (VCO) 72 which provides aclock for the Analogue to Digital Converter (ADC) 36 sampling. The VCO72, ADCs 36, equaliser 74 and phase detector 92 form a Phase Locked Loop(PLL). The phase detector 92 output is also fed forward to digital delayelement 94 which acts to delay the samples output from the equaliser 74.The digital delay element 94 provides a time adjustment function actingon the samples passed to that element. The digital delay element 94 canbe implemented using an interpolator, for example as described in H.Meyr, M. Moeneclaey, S. A. Fechtel, ‘Digital Communication Receivers’,Wiley & Sons, ISBN 0-471-50275-8. Chapter 9, which is incorporatedherein by reference.

The PLL has a relatively low bandwidth of the order of 50-100 kHz andremoves the frequency offset and slowly varying phase offset from theclock. The low bandwidth of the PLL means that the processing delay ofequaliser 74 does not cause significant gain peaking in the feedbackloop, as seen when the bandwidth of the PLL is sufficient to also removethe fast varying phase offset.

The delay element 94 removes higher frequency phase jitter remaining onthe signals by acting as a digital interpolator to delay the position ofthe samples output from equaliser 74. The Low pass filter 95 prior tothe delay element has a bandwidth of the order of 1-4 MHz.

The Low pass filter may be combined with the phase detector, providedthe filter bandwidth is at least about 10 times that of the PLL, suchthat it effectively only affects the feedforward path. Regardless ofwhether the low pass filter is incorporated into the phase detector, oris separate, the bandwidth of the external loop should be lower than thebandwidth of the phase detector and delay element 94.

The combination of a lower frequency feedback loop and a delay elementcontrolled by a feedforward signal mitigates at least some of theproblems described with prior systems and enables a clock recoverysystem to meet the required standards.

FIG. 11 shows the G.8251 Jitter mask and a series of plots for thesystem of FIG. 10. Dashed line 100 marks the minimum jitter values whichmust be tolerated. The simulation assumed a processing delay of 372 ns(125 clock cycles), and a 100 kHz analogue closed-loop bandwidth. Evenwith this relatively large processing delay, the standard can becomfortably met with a Low Pass Filter with a 5.4 MHz 3 dB point. FIG.12 shows a block diagram of a receiver system for correcting distortion,performing clock recovery, and decoding received data. The system shownin FIG. 12 relates to a single polarisation and is duplicated (apartfrom the photodiodes and ADCs which are shared) for the secondpolarisation.

As explained previously, four photodiodes 35 receive four outputs from90° hybrids mixing the optical signal with the optical Local Oscillator.The outputs of those photodiodes are digitised using ADCs 36, thesampling clock of which is provided by VCO 72 configured in conjunctionwith phase detector 92 as described in relation to FIG. 8.

The upper pair of photodiodes (solid signal lines) are for a firstoutput of the polarisation splitter described above and the lower pair(dashed signal lines) for the second output of that splitter. Thesignals from each pair are combined to give a complex valued signal(composing of real and imaginary parts) 110, 111 and passed to theequaliser 112. As explained previously, the ‘butterfly’ equaliserstructure corrects for the unknown polarisation of the incoming opticalsignal.

Delay element 113 acts on the output of the equaliser 112 to remove anyhigh frequency phase jitter not tracked by the PLL. The output of thedelay element 113 is passed to a carrier recovery system 114 to removethe Local Oscillator offset. Example carrier recovery systems aredisclosed in Viterbi, A. (1983), ‘Nonlinear estimation of PSK-modulatedcarrier phase with application to burst digital transmission’,Information Theory, IEEE Transactions on 29(4), 543-551, and H. Meyr, M.Moeneclaey, S. A. Fechtel, ‘Digital Communication Receivers’, Wiley &Sons, ISBN 0-471-50275-8. pp. 311-322, which are incorporated herein byreference.

Decision circuit 115 decides values of the symbols (in this exampleDP-QPSK system the output from the equaliser represents a QPSK signaland therefore each symbol carries two bits) and outputs the decidedsymbols at 119.

A decision-directed tap update system 116 is provided to control andmaintain the tap weights of the equaliser 112 such that the systemcontinuously monitors and tracks the incoming signals. The tap updatesystem 116 operates a tap update algorithm which takes inputs of theinput signal 117, the equalised signal 118 and the decided symbols 119.To ensure correlation between the input signal and the equalised signal118 a second delay element 1100 is utilised on the input signal feed tothe tap update system to mimic the delay applied by the first delayelement. The input signal passed to the update algorithm is thereforeactually a delayed input signal 1101 such that the algorithm does notsee the delay introduced by the first delay element 113. The decidedsymbols signal 119 also differs from the input signal 1101 and equaliseddata 118 as the carrier offset has been removed. The decided symbolsmust therefore be ‘re-spun’ such that they correlate with the othersignals. This is achieved by applying 1102 the output of the carrierphase estimator 1103 to the decided symbols.

The tap update algorithm may be any suitable algorithm for providing therequired functionality, and various options are known in the art. By wayof example, the following description is given in relation to a LeastMean Squares (LMS) algorithm.

Firstly, an error vector of the difference between the equaliser outputand the decided symbol is calculated as below:ē _(k) =y _(k) −p _(k)

We note the use of a line over variables to indicate a complex value,and bold type to indicate a matrix. Where y_(k) is the decided symboloutput:y=d (k)e ^(+j{circumflex over (ø)}) ^(k)

And p_(k) is the equalizer output:p=w ^(H) vwhere H is the Hermitian transpose (or conjugate transpose). The tapcoefficients are then calculated:w _(k+1)=(1−α) w _(k) +μe _(k) *v _(k)

Where w=tap weight matrix, α=leakage factor, μ is the update rate and vis the unequalised input signal matrix. The delayed input signal v _(k)is multiplied by the error value, ē_(k), and by μ, the tap update rate.The first terms in the equation apply the ‘leakage factor’ α to theprevious tap weights, which causes them to decay over time.

The receiver system of FIG. 12 thus provides a system for equalising areceived signal, performing clock recovery, dynamically updating theequaliser to track changing conditions, and determining the value of thereceived symbols. As noted previously, a comparable system may beutilised to process the second polarisation of a DP-QPSK system.

FIG. 13 shows a block diagram of a phase detector suitable for aDP-DQPSK system. The operation of such a phase detector is described inOerder, H. (May 1988), ‘Digital filter and square timing recovery’,Communications, IEEE Transactions on 36(5), 605-612 and Zhu, M. S. M.(November 2005), ‘Feedforward symbol timing recovery technique using twosamples per symbol’, Circuits and Systems I: Regular Papers, IEEETransactions on [Circuits and Systems I: Fundamental Theory andApplications, IEEE Transactions on] 52(11), 2490-2500, incorporatedherein by reference.

For clarity, the following description is given in relation to the firstsignal and blocks shown in FIG. 13, but applies to each of the blocksduplicated in parallel in FIG. 13.

As described in the above references the incoming signal 120 is firstmultiplied by a half-rate sine or cosine clock. In the implementationshown in FIG. 13 which operates using two samples per symbol, this isimplemented by passing alternate samples to separate data paths 121 and122. Path 121 receives the even (A) samples and path 122 receives theodd (B) samples. Even and odd are used as labels only to distinguish onesub set of samples from the other. The samples are then passed throughlow pass filter 123, for example a Finite Impulse Response (FIR) filterto remove high frequency components from the signal which would distortthe phase detector output when working with 2 samples per symbol. Thecomplex value (u+vj) is then squared at block 124 to give a clock phasevector 125.

The four clock phase vector signals are added in adders 120, 121. Adders120, 121 are selectable to allow the selection of one or more of theclock phase vector signals. When in steady state it is likely to be mostpreferable for the phase detector to operate on all signals (i.e. bothpolarisations) to give the best accuracy. However, during startup, or atother times, it is possible that the output of the equaliser for onepolarisation is poor and thus the phase detector may utilise only thegood polarisation to detect the phase. The phase detector may thereforebe switched dynamically based on the system's performance to utilise oneor both of the polarisation signals.

The clock phase vectors are then averaged at blocks 122, 123 andconverted to a phase value using a tan(½π)tan⁻¹ (N/D)) block 124.Averaging the phase vectors rather than the phase value makes the systemmore robust against cycle slips. The parameters of the Average blocks122, 123 define the bandwidth of the phase detector.

Unwrap block 125 removes discontinuities at +/−π radian intervals andallows the phase detector to track phase changes over multiple unitintervals as shown in FIG. 14 (a plot of Phase Detector Output (UnitIntervals (UI)) against Input Phase (UI)) which shows an idealizedoutput. As shown in FIG. 14 the output should be linear since the signalis utilized as a feedforward signal to control Delay Element 113. DelayElement 113 cannot provide an infinite delay and so saturation block 126saturates the signal at a predetermined value (2 unit intervals in FIG.14).

The phase value output is split and may be adjusted 127, 128 accordingto tap weight phase detector 1202 as described below. The outputs 129,1200 are utilised as inputs to the VCO 72 and Delay Element 113,respectively.

FIG. 15 shows a block diagram of a clock frequency offset monitor 1201which may be provided to analyse and indicate whether the clock recoverysystem has locked to the received data signal. Successive phase valuesare compared at block 140 to give the phase change between those values.Block 141 performs an unwrap function to prevent false indications whenthe phase moves over an edge of the tan θ function. Infinite ImpulseResponse (11R) filter 142 filters the signal resulting in a signal 143indicating the offset between the received data clock and the localclock. When this value is stable, lock has been obtained. The indicationof clock lock may be utilised by the processing system to controldecoding of data or for general system control.

FIR filters, such as those used in the equaliser, can act as variabledelay elements by interpolating between samples. This occurs by the tapweights shifting to the left or the right. FIGS. 16a and 16b show plotsof tap weights of an FIR filter, each having the same impulse responsebut the tap weights in FIG. 16b are shifted off-centre to introduce adelay compared to the weights in FIG. 16a . The centred weights shown inFIG. 16a are better able to correct for increased distortion than theoff-centre ones shown in FIG. 16b , and therefore it is preferable forthe tap weights to be centralised for optimum equaliser performance.

The equaliser 112, being an FIR, attempts to track and correct any slowclock phase drift which causes the tap weights to move from theircentral position. The off-centre tap weights are less able to equaliseincreased distortion and therefore system performance may be degraded.The slow clock phase drift should be corrected by the PLL, not theequaliser.

A tap weight phase detector 1202 is provided to monitor thecentralisation of the tap weights and provide a correction signalindicative of the tap weight centre offset.

The correction signal output by tap weight phase detector 1202 is splitfor use by the VCO 72 and delay element 113. The signal for the VCO 72passes to multiplier 1203 where it is multiplied by a coefficientK_(tapPLL). The signal for the delay element passes to multiplier 1204where it is multiplied by a coefficient K_(tapDE). K_(tapPLL) andK_(tapDE) define the magnitude of the correction signal that is added tothe phase value for the VCO 72 and Delay Element 113 respectively.Adders 127 and 128 add the correction signal to the phase signal for theVCO 72 and delay element 133 respectively.

In a first example, K_(tapDE) is set to zero such that the correctionsignal is only applied to the VCO 72. In a second example, K_(tapDE) maybe non-zero such that the correction is applied to both the VCO 72 andDelay Element 133.

The correction signal applied to the VCO 72 (and Delay Element 113 ifK_(tapDE)≢0) causes the tap weights to re-centralise under the action ofthe tap update system.

In an alternative implementation, the correction signal could beutilised to directly adjust the tap weights. The signal should beapplied slowly to allow the VCO 72 and Delay Elements 133 to track thechange thereby avoiding degradation of the equaliser performance.

The tap weight centre for the x-polarisation can be calculated using thefollowing equation:—

${\hat{n}}_{x} = \frac{\sum\limits_{n}{n( | {\overset{\_}{h}}_{{xx}_{n}} \middle| {}_{2}{+ | {\overset{\_}{h}}_{{yx}_{n}} |^{2}}  )}}{\sum\limits_{n}( | {\overset{\_}{h}}_{{xx}_{n}} \middle| {}_{2}{+ | {\overset{\_}{h}}_{{yx}_{n}} |^{2}}  )}$

Where h _(xx) _(n) and h _(yx) _(n) are the nth elements of thecomplex-valued sub matrices h _(xx) and h _(yx) for the X-X and Y-Xfilters respectively which equalise the X-polarisation:

$\overset{\_}{w} = \begin{pmatrix}{\overset{\_}{h}}_{xx} \\{\overset{\_}{h}}_{yx}\end{pmatrix}$

A comparable equation is used for the Y-polarisation. The tap weightphase (the difference between the tap weight centre and the central tap)is then given by:φ_(tapX) =n _(centre) −{circumflex over (n)} _(X)

Where n_(centre) is the number of the central tap.

The tap weight phase is combined with the output of the phase detector,as shown in FIG. 13, to correct the phase of the VCO 72 and DelayElement 113, 1100, which in turn results in the tap weights moving tothe centralised position. The tap weight phase signal output may be acombination of both the X and Y polarisations, or may be based on one orthe other polarisation. It may be desirable to select from one or bothpolarisations depending on the relatively quality of the signals, or ifone of the polarisations has not correctly acquired stable tap weights.

On startup of a transmission system the receiver has no knowledge of thecorrect clock phase or tap weights for the equaliser. For the phasedetector to acquire the phase of the signal (and hence for the clockrecovery system to operated) the equaliser must, at least to a certaindegree, equalise the received signal. However, determination of the tapweights for the equaliser relies on the clock recovery system.

FIG. 17 shows a flow chart of a system for acquiring the clock phase andinitial tap weights to allow startup of the system. FIG. 19 provides analternative view of the methods of FIGS. 17 and 18 to exhibit thefunctions performed by each part of the system and the transfer of databetween the ASIC and the DSP during the acquisition 187 and tracking 188phases. Reference numerals on FIG. 19 relate to the blocks of FIGS. 17and 18.

At block 160 a series of samples 180 of the data 183 is acquired by theASIC 181. For example, 4000 samples may be acquired, which at a typicalsample rate of two samples per symbol, gives 2000 symbols. At block 161the series of samples is transferred to a Digital Signal Processor (DSP)182 for processing according to the programming of that DSP. At block162 the DSP applies a blind-optimisation algorithm to the samples 180 inorder to identify an initial set of tap weights which may be used toequalise the incoming signal.

The use of a DSP 182 associated with the ASIC 181 for performing thealgorithm is convenient as it may be programmed to perform the specificalgorithms required. Since the algorithm need not be applied in realtime there is a reduced requirement on the performance of the DSP 182compared to the processing system which processes the received data inreal time. Providing this function in the ASIC 181 is likely to besubstantially more expensive and complex than utilising a DSP 182 andtherefore the use of two devices may be more cost effective. However,the method can equally be applied within a single device if appropriate.In addition, ASIC implementations will typically work in a highlyparallel manner, and the update rate μ may be limited by feedbackdelays. A non-real time implementation allows substantially higherupdate coefficients to be used, and the same data may be processedseveral times until the tap weights have converged. The use of a shortblock of data means that clock offsets (which may be 100-200 ppm) arenot significant to prevent acquisition of an initial set of tap weights.

At block 163 the initial tap weights are transferred to the ASIC 181 andapplied to the equaliser. The equaliser output is now a partiallycorrected signal from which the phase detector can operate. At block 164the PLL is activated. Provided the channel is stable, the output of theequaliser will be valid and the PLL will acquire the clock at block 165and will remove the clock offset.

Once the clock has been acquired, at block 166 the digital delayelements and decision-directed feedback systems are activated and thesystem enters tracking mode. Provided the initial tap estimates aresufficiently accurate the system will optimise the tap weights, PLL anddelay elements. In order for this to succeed the tap weights must besufficiently accurate for the decision directed update to function,which requires approximately <˜10⁻²BER.

Algorithms for the blind-compensation of DP-QPSK are known, for exampleas described in Raheli, R. & Picchi, G. (1991), Synchronous andfractionally-spaced blind equalization in dually-polarized digital radiolinks, in ‘Communications, 1991. ICC 91, Conference Record. IEEEInternational Conference on’, pp. 156-161 vol. 1, incorporated herein byreference. A further known blind acquisition method which may beapplicable in relation to the method of FIG. 17 is a constant modulusalgorithm as described in Godard, D. (1980), ‘Self-RecoveringEqualization and Carrier Tracking in Two-Dimensional Data CommunicationSystems’, Communications, IEEE Transactions on [legacy, pre-1988]28(11), 1867-1875, incorporated herein by reference.

The optimisation algorithm utilised in block 162 may be applied a numberof times to the series of samples. A particularly efficient method is toapply the algorithm to the samples alternately forwards and backwardssuch that any clock offset is not relevant. The update coefficients usedin the algorithm may be considerable higher than are utilised in acontinual optimisation system as it is only desired to acquire aninitial set of tap weights, not to provide continual performance. Highupdate coefficients tend to lead to sporadic changes in tap weights andhence degradation in performance, but provide a more rapid convergenceto the initial tap weights.

Prior to the tap weights being transferred to the ASIC they may bere-centred 184 by shifting the taps in response to a calculation of thetap weight centre, or by applying a shift using a digital interpolationfunction.

The method shown in FIG. 17 assumes that the acquisition phase,polarisation or channel model does not change significantly during theperiod of the acquisition. It is reasonable to assume that thepolarisation and channel model does not change during this phase(approximately 20-500 ms), and a maximum clock offset of 250 ppm isreasonable. With that offset a drift of 0.5 unit intervals will occur inthe 2000 symbols, which equates to a shift of 1 equaliser tap. This isnot a significant shift and so the required assumptions are reasonable.If the assumptions are not met, then a new series of samples may beacquired and the process restarted.

FIG. 18 shows a flow chart of an extension of the method of FIG. 17 toimprove the equalisation before tracking mode is activated. After theclock has been acquired at block 165 a further series of samples 185 areacquired at block 170. At block 171 those samples 185 are transferred tothe DSP 182 and again processed using the blind optimisation algorithmat block 172. Since the clock offset has been removed from this secondseries of samples a more accurate set of tap weights can be derived. Atblock 173 those tap weights (after optional recentering 186) aretransferred to the equaliser and applied to the received signal.

At block 174 the decision-directed tracking and digital delay elementsare activated and the receiver enters tracking mode. Frame detection maythen be obtained and the system goes on to continuously track thechannel.

All parameters and results given in the above description relate to a 40Gbit/s DP-QPSK transmission system and are given to describe thatsystem. All parameters may be modified in conventional manners asrequired based on the particular system being utilised without departingfrom the invention.

Where functions or algorithms have been described as being performed bya particular device or type of device, this is for example only and isnot intended to be limiting in any way. As will be appreciated anysuitable mode of implementation may be utilised as appropriate.

The blocks and demarcation between functions described above is given byway of example only, and as will be appreciated the functions may bedemarked and distributed in any suitable way. Where a function has beendescribed as being performed by a particular type of device (for examplean ASIC) it will be appreciated that the principles described herein arealso applicable to other methods of implementing those functions (forexample in software running on a DSP).

The term Voltage Controlled Oscillator is used to describe a componentwhich produces an output signal for clocking the ADCs in dependence onan input signal. As will be appreciated, it is not intended to restrictthe component to one which is controlled directly by a varying voltage,but rather to describe the function of the component. For example, thecomponent may equally be controlled by a digital signal indicatingvalues.

Although not shown or described explicitly, it will be appreciated that,as is well known in the art for PLLs, a loop filter will be incorporatedat a convenient location in the various PLLs described herein. Thatfilter may be conveniently located in close proximity to the clocksource, and may form part of the same device as that source.Alternatively, the filter may be located at any convenient location andprovided by any convenient means.

The manner of implementation of the techniques described herein isdependent on the particular system and the implementation is within thecapabilities of the skilled reader once they have been made aware of thefunctions required by this document.

The applicant hereby discloses in isolation each individual featuredescribed herein and any combination of two or more such features, tothe extent that such features or combinations are capable of beingcarried out based on the present specification as a whole in the lightof the common general knowledge of a person skilled in the art,irrespective of whether such features or combinations of features solveany problems disclosed herein, and without limitation to the scope ofthe claims. The applicant indicates that aspects of the presentinvention may consist of any such individual feature or combination offeatures. In view of the foregoing description it will be evident to aperson skilled in the art that various modifications may be made withinthe scope of the invention.

Any range or device value given herein may be extended or alteredwithout losing the effect sought, as will be apparent to the skilledperson.

It will be understood that the benefits and advantages described abovemay relate to one embodiment or may relate to several embodiments. Theembodiments are not limited to those that solve any or all of the statedproblems or those that have any or all of the stated benefits andadvantages.

Any reference to an item refers to one or more of those items. The term‘comprising’ is used herein to mean including the method blocks orelements identified, but that such blocks or elements do not comprise anexclusive list and a method or apparatus may contain additional blocksor elements.

The steps of the methods described herein may be carried out in anysuitable order, or simultaneously where appropriate. Additionally,individual blocks may be deleted from any of the methods withoutdeparting from the spirit and scope of the subject matter describedherein. Aspects of any of the examples described above may be combinedwith aspects of any of the other examples described to form furtherexamples without losing the effect sought.

It will be understood that the above description of a preferredembodiment is given by way of example only and that variousmodifications may be made by those skilled in the art. Although variousembodiments have been described above with a certain degree ofparticularity, or with reference to one or more individual embodiments,those skilled in the art could make numerous alterations to thedisclosed embodiments without departing from the spirit or scope of thisinvention.

The term ‘computer’ is used herein to refer to any device withprocessing capability such that it can execute instructions. Thoseskilled in the art will realize that such processing capabilities areincorporated into many different devices and therefore the term‘computer’ includes PCs, servers, mobile telephones, personal digitalassistants and many other devices. Similarly DSP or ASIC is not intendedto restrict the invention to any particular type of processing device,but those terms are simply used to refer to one possible implementation.

The methods described herein may be performed by software in machinereadable form on a tangible storage medium. The software can be suitablefor execution on a parallel processor or a serial processor such thatthe method steps may be carried out in any suitable order, orsubstantially simultaneously.

This acknowledges that software can be a valuable, separately tradablecommodity. It is intended to encompass software, which runs on orcontrols “dumb” or standard hardware, to carry out the desiredfunctions. It is also intended to encompass software which “describes”or defines the configuration of hardware, such as HDL (hardwaredescription language) software, as is used for designing silicon chips,or for configuring universal programmable chips, to carry out desiredfunctions.

Those skilled in the art will realize that storage devices utilized tostore program instructions can be distributed across a network. Forexample, a remote computer may store an example of the process describedas software. A local or terminal computer may access the remote computerand download a part or all of the software to run the program.Alternatively, the local computer may download pieces of the software asneeded, or execute some software instructions at the local terminal andsome at the remote computer (or computer network). Those skilled in theart will also realize that by utilizing conventional techniques known tothose skilled in the art that all, or a portion of the softwareinstructions may be carried out by a dedicated circuit, such as a DSP,programmable logic array, or the like.

What is claimed is:
 1. A method of initially acquiring tap weights for afinite impulse response (FIR) filter used to correct distortion in anoptical communication receiver, the method comprising: acquiring andstoring a series of samples of a received optical signal, the series ofsamples being acquired in an order in time; applying a blindoptimization algorithm to the series of samples to obtain firstestimated tap weights for the FIR filter in order to equalize thereceived optical signal, wherein the blind optimization algorithm isapplied to the series of samples alternately in the order that theseries of samples is acquired and in a reverse order that the series ofsamples is acquired in a digital processing system; activating a clockrecovery system in the optical communication receiver based on the firstestimated tap weights to remove a clock offset for subsequent samples;after the clock recovery system is activated, acquiring and storing afurther series of samples of the received optical signal, wherein theclock offset is removed from the further series of samples; and applyingthe blind optimization algorithm to the further series of samples toobtain second estimated tap weights having an accuracy higher than thefirst estimated tap weights.
 2. The method according to claim 1, furthercomprising transferring the second estimated tap weights to the FIRfilter.
 3. The method according to claim 2, further comprisingcorrecting distortion in the received optical signal with the FIRfilter.
 4. The method according to claim 1, further comprising applyingthe second estimated tap weights to the FIR filter and correctingdistortion in the received optical signal based on the second estimatedtap weights.
 5. The method according to claim 1, further comprisingactivating a decision system in the optical communication receiver todecode values represented by the received optical signal.
 6. The methodaccording to claim 5, further comprising entering a tracking mode, inwhich a tap update algorithm is activated and configured to update thesecond estimated tap weights.
 7. An apparatus comprising: an opticalreceiver configured to acquire and store a series of samples of areceived optical signal, the series of samples being acquired in anorder in time; a clock recovery system; a digital signal processorconfigured to: apply a blind optimization algorithm to the series ofsamples to obtain first estimated tap weights, wherein the digitalsignal processor applies the blind optimization algorithm to the seriesof samples alternately in the order that the series of samples isacquired and in a reverse order that the series of samples is acquired;wherein the optical receiver is further configured to activate the clockrecovery system based on the first estimated tap weights to remove aclock offset for subsequent samples; wherein, after the clock recoverysystem is activated, the optical receiver is configured to acquire andstore a further series of samples of the received optical signal and theclock offset is removed from the further series of samples; wherein thedigital signal processor is further configured to apply the blindoptimization algorithm to the further series of samples to obtain secondestimated tap weights having an accuracy higher than the first estimatedtap weights; and a finite impulse response (FIR) filter configured toequalize the received optical signal with the second estimated tapweights.
 8. The apparatus of claim 7, wherein the digital signalprocessor is further configured to transfer the second estimated tapweights to the FIR filter.
 9. The apparatus of claim 7, wherein the FIRfilter is further configured to apply the second estimated tap weightsand correct distortion in the received optical signal based on thesecond estimated tap weights.
 10. The apparatus of claim 7, wherein theoptical receiver is further configured to activate a decision system todecode values represented by the received optical signal.
 11. One ormore non-transitory computer readable storage media encoded withcomputer readable instructions which, when executed by a processor,cause the processor to: acquire and store a series of samples of areceived optical signal, the series of samples being acquired in anorder in time; apply a blind optimization algorithm to the series ofsamples to obtain first estimated tap weights for a finite impulseresponse (FIR) filter in order to equalize the received optical signal,wherein the blind optimization algorithm is applied to the series ofsamples alternately in the order that the series of samples is acquiredand in a reverse order that the series of samples is acquired in adigital processing system; activate a clock recovery system in anoptical communication receiver based on the first estimated tap weightsto remove a clock offset for subsequent samples; after the clockrecovery system is activated, acquire and store a further series ofsamples of the received optical signal, wherein the clock offset isremoved from the further series of samples; and apply the blindoptimization algorithm to the further series of samples to obtain secondestimated tap weights having an accuracy higher than the first estimatedtap weights.
 12. The one or more non-transitory computer readablestorage media of claim 11, wherein the instructions further cause theprocessor to correct distortion in the received optical signal with theFIR filter.
 13. The method of claim 1, further comprising: shifting thesecond estimated tap weights toward a tap weight center.
 14. The methodaccording to claim 13, wherein the shifting is performed using a digitalinterpolation function.
 15. The apparatus of claim 7, wherein thedigital signal processor is further configured to shift the secondestimated tap weights toward a tap weight center.
 16. The apparatus ofclaim 15, wherein the digital signal processor is configured to shiftthe second estimated tap weights using a digital interpolation function.17. The apparatus of claim 10, wherein the optical receiver isconfigured to enter a tracking mode in which a tap update algorithm isactivated and configured to update the second estimated tap weights. 18.The one or more non-transitory computer readable storage media of claim11, wherein the instructions further cause the processor to shift thesecond estimated tap weights toward a tap weight center.
 19. The one ormore non-transitory computer readable storage media of claim 18, whereinthe second estimated tap weights are shifted using a digitalinterpolation function.
 20. The one or more non-transitory computerreadable storage media of claim 11, wherein the instructions furthercause the processor to enter a tracking mode and activate a tap updatealgorithm configured to update the second estimated tap weights.